Method and device for estimating transmitted signals in a receiver in digital signal transmission operations

ABSTRACT

In a digital signal transmission system, a receiver receives a signal, wherein the signal bandwidth of the system exceeds the system symbol rate. A correlation and sampling circuit receives a baseband signal, samples the signal eight times per symbol time, correlates, generates a channel estimate and down-samples the sampled signal to form an observed signal. This signal is filtered in a prefilter, whose output is sampled at symbol rate and the obtained signal is delivered to a channel equalizer which performs a viterbi algorithm with non-quadratic metric calculation and generates estimated symbols. A channel estimation filter receives a symbol sequence which contains alternate zero-value symbols and the estimated symbols and generates an estimated signal. An error signal is generated and used to adapt the channel estimate and also to generate weight factors. The coefficients of the prefilter are generated as a function of the channel estimate and the weight factors. Coefficients are generated in a metric calculation filter, by convolving the channel estimate with the prefilter and are used to generate the estimated symbols. The transmission channel, excluding the prefilter, is estimated explicitly so as to enable fast channel changes to be followed. The use of the weight factors enables a short channel estimate to be used. The insertion of the zero-value symbols simplifies adaptation of the channel estimate.

BACKGROUND

The present invention relates to a method in digital signal transmissionover a channel for estimating in a receiver transmitted symbols from atransmitted radio signal, wherein the symbol estimation is effected in achannel equalizer in accordance with a selected viterbi algorithm, andwherein the method comprises the following method steps:

receiving and demodulating the transmitted signal to a received signal;

sampling the received signal at at least one sampling time point persymbol;

determining at least an initial value of the estimated channel impulseresponse, the channel estimate, with the aid of the sampled signalvalues;

determining a symbol sampling time point;

selecting filter coefficients of a prefilter and filtering the sampledsignal values in the prefilter to obtain prefiltered, observed signalvalues; and

generating at least preliminarily estimated symbols in accordance withthe viterbi algorithm, with the aid of the prefiltered, observed signalvalues.

The invention also relates to an arrangement for carrying out themethod.

One problem which often occurs in the digital radio transmission of asignal over a channel is that a transmitted signal is subjected to noiseand co-channel disturbance and also to multipath propagation whichresults in time dispersion. For instance, in the case of mobiletelephony, the transmission properties of the radio channel shift as aresult of the transmitter and receiver changing their mutual relativepositions. These problems have been solved in time-shared digital radiotransmission systems in that the signal sequences that are transmittedin a time slot have one or more synchronizing sequences and one or moredata sequences. The synchronizing sequence is known to the receiver andthe receiver is able to estimate the transmission properties of thechannel, i.e. to make a channel estimate, with the aid of this sequence.The receiver estimates the symbols of the data sequence containing theinformation that is to be transmitted, with the aid of this channelestimate.

In certain cases, it is not sufficient to make a channel estimate onlyonce with each time slot. In the case of long time slots, thetransmitter and the receiver have time to change their mutual relativepositions considerably within the duration of the time slot. This meansthat the transmission properties of the channel can also changeconsiderably within the duration of the time slot, so that theestimation of the transmitted symbols made by the receiver will bedeficient and the transmitted information therefore unclear orambiguous. A radio receiver in which such disturbances are partiallyavoided is described in an article in IEEE Transactions on InformationTheory, January 1973, pages 120-124, F. R. Magee, Jr. and J. G. Proakis:"Adaptive Maximum-Likelihood Sequence Estimation for Digital Signalingin the presence of Intersymbol Interference". The article describes achannel equalizer which includes a viterbi analyzer having an adaptivefilter as a channel estimating circuit. Received symbols are comparedsuccessively with hypothetical symbols and those hypothetical symbolswhich coincide closest with the received symbols are selectedsuccessively to form an estimated symbol sequence. The parameters of theadaptive filter are adjusted successively to the changed channel, withthe aid of the selected, decided symbols.

A description of the viterbi algorithm is given in an article by G.David Forney, Jr.: "The Viterbi Algorithm" in Proceedings of the IEEE,Vol. 61, No. 3, March 1973. The article also describes in some detailthe state and state transitions of the viterbi algorithm and also howthese state transitions are chosen so as to obtain the most probablesequence of symbols.

The signal transmission between transmitter and receiver may beconnected with certain problems, despite performing sequence estimationand adaptive channel estimation in the aforedescribed manner. One reasonfor these deficiencies is that the signal bandwidth of the systemexceeds the system symbol rate, so-called excess bandwidth, as is thecase, for instance, in the North American mobile telephone system TIAIS-54. Another reason for these deficiencies is that the transmissionproperties of the channel can change quickly, for instance as a resultof fading. Two different types of solution to the problem of symbol rateare known to the art, in which a MLSE-detector (Maximum-LikelihoodSequence Estimator) is used:

The viterbi algorithm itself operates at a higher rate than the symbolrate.

An adaptive, fractionally spaced prefilter is used prior to the viterbianalyzer.

The first type of solution is described in an article by Yongbing Wan,et al, of NovAtel Communications Ltd.: "A Fractionally-SpacedMaximum-Likelihood Sequence Estimation Receiver in a Multipath FadingEnvironment" published in the Proceedings of IEEE, ICASSP 1992.According to this article, a received radio signal is sampled twice witheach symbol and the channel estimation is effected with the aid of anadaptive filter that uses this double sampling rate. The symbol estimateis performed in a viterbi analyzer which also uses the double samplingrate. The delta metric values, i.e. deviations between the received andthe hypothetical sequences, are calculated for both the samplingoccasions per symbol and the two delta metric values are added todetermine a best state transition according to the viterbi algorithm.When adapting the filter with the aid of the estimated symbols, afictive symbol is inserted at each alternate sampling time point. Thesefictive symbols are produced by interpolation between the estimatedsymbols in a second filter. The proposed solution has certain drawbacks.It is necessary to sample the received symbols at highly specific timepoints, and the adaptive channel estimation is complex. Theinterpolation in the second filter results in delays which may impairthe symbol estimation. Filters that are used in signal processing, forinstance a transmitter filter or a receiver filter must be knownfilters. Receiver filters, which may contain coils and capacitors, causeparticular problems due to aging, manufacturing accuracies andtemperature variations.

Another solution of the first kind is given in a paper written by R. A.Iltis: "A Bayesian Maximum-Likelihood Sequence Estimation Algorithm forA-Priori Unknown Channel and Symbol Timing" Department of Electrical andComputer Engineering, University of California, Santa Barbara, Aug. 21,1990. This paper also states that a received signal shall be sampledtwice with each symbol. Symbol estimation is effected in accordance witha viterbi algorithm, which calculates two delta metric values for eachsymbol, and these two values are weighted in the metric calculation. Thechannel estimation is performed in an adaptive filter having filtercoefficients of the spacing of a symbol time, although the coefficientsare adapted with each sampling occasion, thus twice with each symbol.The solution given includes a comprehensive metric calculation andbecause the channel estimate used has its filter taps at a full symboltime spacing, it fails to solve the problem of symbol synchronization inrespect of complicated rapidly varying excess bandwidth channels. Also,similar to the solution proposed by Yongbing Wan according to theaforegoing, a receiver filter must be known with high degree of accuracyin the receiver.

The aforesaid methods relating to the first type of problem solution forsolving the problem of low symbol rate are relatively demanding withregard to the calculations that must be carried out. A method whichrelates to the second type of solution has been proposed in an articlein IEEE Transactions on Communications, Vol. Com-22, No. 5, May 1974,written by G. Ungerboeck: "Adaptive Maximum-Likelihood Receiver forCarrier-Modulated Data-Transmission Systems". According to this article,a received radio signal is sampled several times with each symbol timeand the sample signal is allowed to pass through a prefilter. Theprefiltered signal is sampled down to symbol rate and is then processedin a viterbi analyzer, which produces estimated symbols. The sampledimpulse response of the radio channel is estimated with the aid of achannel estimate, this response including both the actual channelbetween transmitter and receiver, transmitter filter and receiver filterand also the prefilter. The prefilter and the channel estimation filterare each adapted to the variable radio channel with the aid of theestimated symbols obtained from the viterbi analyzer. This analyzer usesthe filter coefficients in the channel estimation filter to performsymbol estimation, in a known manner. The metric calculation in theviterbi analyzer is non-quadratic and is simplified in comparison withthe quadratic metric calculation normally used. This non-quadraticmetric calculation can be used because the received signal has beenfiltered in the prefilter. The simplified method defined in theUngerboeck article requires certain restrictions in the adaptationalgorithm, as illustrated in an article in Proceedings of the IEEE, Vol.73, No. 9, September 1985, pages 1370-1372 by S.U.H. Qureshi: "AdaptiveEqualization". The restrictions are required because the channelestimation filter and the prefilter are each separately adapted with theaid of the estimated symbols. This can result in all of the coefficientsin the two filters converging towards zero. The restrictions areintroduced with the intention of counteracting this convergence, theserestrictions, for instance, comprising assigning a fixed value to one ofthe coefficients in the channel estimation filter. On the other hand,these restrictions render the simplified method less suited for use withfast varying channels, for instance fast fading channels. The problemthat occurs resides in the lack of time in which to achieve thisadaptation and, in principle, the same sort of problem of following thechannel behaviour occurs as that occurring in a linear or aDFE-equalizer (Decision Feed Back). Expressed in simple terms, thismeans that an attempt is made to follow the inverted impulse response ofthe channel rather than the actual channel impulse response itself, andit is well known that the channel has, in general, a much slowerchanging rate than its inverse.

The second of the aforesaid problems, the fading problem, has earlierbeen solved, as described for instance in Swedish Patent Application SE9102612-0, which corresponds to U.S. patent application Ser. No.07/942,270 filed Sep. 9, 1992. A complex value signal is transmittedbetween transmitter and receiver and the signal strength of the signalvaries very quickly and has abrupt fading dips. According to this patentapplication, it is observed that the real and imaginary components ofthe signal each vary relatively regularly and that the time derivativesof these components are often almost linear. This is utilized toestimate both the radio channel impulse response and the derivative ofthe impulse response. This derivative estimate is used to estimate theimpulse response after a fading dip, during which the radio signal hasbeen practically extinguished. A similar method is described in adissertation by Lars Lindblom: "Adaptive Equalization for Fading MobileRadio Channels", System and Control Group, Department of Technology,Uppsala University, 1992.

SUMMARY

The present invention relates to a method and to an arrangement forsymbol estimation in digital signal transmission over a channel. Thoseproblems which occur when the digital transmission system has a signalbandwidth which exceeds the symbol rate of the system, referred to as"excess bandwidth", are solved by one aspect of the invention. Thoseproblems which occur in the case of rapidly varying channels, forinstance rapidly fading radio channels, is solved by another aspect ofthe invention.

The method is effected with the aid of a viterbi analyzer which utilizesthe non-quadratic metric calculation according to the aforesaid articleby G. Ungerboeck. A received signal is sampled several times with eachsymbol time and this observed, sampled signal is permitted to passthrough a fractionally sampled prefilter. The prefiltered signal issampled down to symbol rate and is applied to the viterbi analyzer. Theviterbi analyzer performs the symbol estimation at symbol rate andproduces estimated symbols. A system impulse response, includingtransmission channel and transmitter filter and receiver filter, butexcluding the prefilter, is estimated explicitly, such as a channelestimate in a channel estimation filter intended herefor. This filter isoperative in generating estimated values of the received signal with theaid of the estimated symbols obtained from the viterbi analyzer. Errorsignals are formed as a difference between the estimated signals and theobserved, sampled signals. The prefilter is generated essentially as atime-inverted and complex-conjugated version of the channel estimate.However, the prefilter coefficients are weighted with the aid of weightfactors which, according to one advantageous embodiment of theinvention, are generated in dependence on the inverted values of theerror signals. This enables a short channel estimate containingrelatively few coefficients to be used. As before mentioned, theprefilter is fractionally sampled and consequently it is also necessaryto fractionally sample the channel estimate. In order to enable thesimplified metric calculation to be carried out in the viterbi analyzer,coefficients in a metric calculating filter are generated. This filterincludes generally a convolution between the channel estimate and theprefilter and is thereafter sampled at symbol rate.

As before mentioned, the estimated values of the received symbols andthe error signal are generated with the aid of the determined symbolsobtained from the viterbi analyzer. According to the aforegoing, thesymbols received upstream of the prefilter are sampled several timeswith each symbol time and fictive symbols are inserted into the sequenceof determined symbols at these intermediate sampling time points, toenable the error signals to be generated at the sampling time pointsbetween the symbols. The fictive symbols are assigned zero-values. Thismeans that all transmitter and receiver filters will be included in thechannel estimate and that these filters need not therefore be known tothe receiver. Another result of introducing the zero-value symbols isthat the channel estimation becomes much less complex.

It is often necessary to adapt the estimated impulse response of thechannel, the channel estimate, for instance in the case of long timeslots or a rapidly changing channel. The error signals and anappropriate adaptation algorithm are used in this regard. Thisadaptation, which is made several times with each symbol time, isconsiderably simplified by inserting the zero-value symbols. In thisway, new coefficient values in the filter need only be generated oncewith each symbol, irrespective of the number of sampling time points persymbol. By inserting the fictive symbols of zero-value instead of usingfurther interpolation filters to generate the fictive symbols, it isalso possible to use a relatively short channel estimate. This resultsin relatively short delays when adapting the channel estimate, whichalso contributes to enabling the symbols to be estimated with a highdegree of accuracy in the viterbi analyzer.

A predictor can be used to advantage in an adaptive viterbi detectorwhen generating the prefilter and the metric calculation filter. Thisenables some compensation to be made for the delayed channel estimatethat results from the decision delay in the viterbi analyzer. Accordingto one embodiment of the present invention, different coefficients ofthe prefilter are also predicted for different future time periods. Thisis advantageous when a non-quadratic viterbi analyzer having a prefilteris used.

In those instances when only the problem involving the rapidly varyingchannel is to be solved, the second aspect of the invention, thereceived signal need only be sampled once with each symbol. In thisregard, the channel estimate, and also the prefilter and metriccalculation filter, are sampled only once with each symbol. As in theaforegoing, the viterbi analyzer works at symbol rate.

It can be said in summary that the present invention distinguishes fromthe known techniques essentially in the three following respects:

The channel is estimated and tracked explicitly and the coefficients inthe prefilter and the metric calculation filter are calculated with theaid of the channel estimate obtained.

The weighting factors are used when generating the prefilter and themetric calculation filter to enable a short channel estimate to be used,with litle loss in performance.

Prediction times of different lengths are permitted for the coefficientsin the prefilter.

The invention can be applied generally in signal transmission andparticularly for rapidly fading radio channels. The receiver performanceis improved considerably in comparison with known techniques, withoutthe receiver needing to be too complex.

BRIEF DESCRIPTION OF THE DRAWINGS

An exemplifying embodiment of the invention will now be described inmore detail with reference to the accompanying drawings, in which

FIG. 1 is a block schematic outlining a transmitter and a receiver in adigital radio system;

FIG. 2 illustrates time slots and a symbol sequence for time-sharedradio transmission;

FIG. 3 illustrates a complex number plan with symbol values;

FIG. 4 is a block schematic illustrating the receiver;

FIG. 5 is a block schematic illustrating a channel estimation filter;

FIG. 6 is a diagrammatic illustration of a radio channel impulseresponse;

FIG. 7 is a block schematic illustrating a prefilter;

FIG. 8 is a block schematic illustrating a weighting factor generatingcircuit;

FIG. 9 is a flowsheet illustrating the inventive method;

FIG. 10 is a block schematic illustrating an alternative embodiment ofthe invention; and

FIG. 11 is a block schematic illustrating a further embodiment of theinvention.

DETAILED DESCRIPTION

FIG. 1 illustrates schematically a radio transmission system fortime-shared signal transmission. A transmitter includes a unit 10 whichreceives an information carrying signal and generates correspondingdigital symbols S(k). In the reference S(k), the letter k is an integersymbol counter. These symbols are delivered to a unit 11 which includesa transmitter filter and a digital/analogue converter. The symbols S(k)are signal processed in the unit 11 and transmitted to a radiotransmitter 12, which transmits the signal made analogue in the unit 11in the form of a signal R(T) of selected carrier frequency. Theanalogued signal is transmitted over a radio channel 13 to a receiverequipped with a radio receiver 14. The channel 13 subjects the signalR(T) to multipath propagation among other things, as indicated in theFigure by double signal paths. The signals travelling along one signalpath are reflected against, for instance, a building 18 prior toreaching the receiver. The radio receiver 14 demodulates the receivedsignal to baseband and delivers a baseband signal y(T) to a correlatingand sampling circuit 15. In turn, the circuit delivers an observedsampled signal referenced y(2). The signal y(k/2) is received by aprefilter circuit 20 which delivers a prefiltered, observed signal z(k)to a channel equalizer 17. The signal z(k) is processed in the channelequalizer 17 in accordance with a viterbi algorithm and the equalizerdelivers estimated symbols S_(D) (k), which shall coincide as near aspossible to the symbols S(k) of the transmitter. The viterbi algorithmuses a simplified non-quadratic metric calculation according to theaforesaid reference G. Ungerboeck. The correlating and sampling circuit15 is connected to a channel estimation circuit 16 and delivers theretothe initial values of a channel estimate which includes the channel 13.According to this embodiment, the circuit 16 is adaptive and generatessuccessively new coefficient values for the channel estimate, which ishereby adapted successively to the time-varying channel 13 with the aidof the signal y(k/2) and the estimated symbols S_(D) (k). In addition tothe channel estimate, there are also generated in the channel estimationcircuit 16 the coefficient values of the prefilter circuit 20 and thecoefficient values of a metric calculation filter that is used by theviterbi algorithm in the channel equalizer 17 when estimating thesymbols S_(D) (k). It can be said generally that it is the filters inthe channel estimation circuit 16 and the generation of its filtercoefficients that are the subject of the invention, as will be describedin more detail further on.

It should be noted that, for instance, the estimated symbols S_(D) (k)are delayed in relation to the observed signals z(k), despite the factthat the same symbol counter (k) is given. This reference method is usedthroughout the following description for the sake of simplicity and itwill be understood that the person skilled in this art will realize thatcertain circuits cause delays. It has only been considered necessary tostate explicitly a delay of a signal in a few instances. The referencesign (k-1) thus denotes a delay comprising one symbol time.

As before mentioned, the radio transmission system according to thisembodiment is time-shared, as illustrated in FIG. 2, in which T denotestime. A carrier frequency, or actually a frequency-pair fortwo-directional communication, is divided into three time slots 19,numbered 1, 2 and 3. A symbol sequence SS which includes a synchronizingsequence SY and two data sequences SD1 and SD2 containing theinformation to be transmitted is transmitted in each time slot. Thesymbol sequence SS includes binary signals, although the aforesaidsymbols S(k) are modulated in accordance, for instance, withQPSK-modulation, as illustrated in FIG. 3. In a complex number planehaving coordinate axes designated I and Q, the four possible values S₀,S₁, S₂ and S₃ of the symbols S(k) are marked, as are also correspondingbinary numbers 00, 01, 10 and 11. The time taken to transmit one suchmodulated symbol is designated one symbol time TS, as schematicallyshown in FIG. 2. It is these whole symbol times TS that are counted bythe integer symbol counter k. The aforedescribed division into timeslots and symbol modulation are known techniques and do not form anypart of the inventive concept.

The system outlined in FIGS. 1 and 2 may exist in a mobile telephonesystem, in which the transmitter is a base station and the receiver is amobile station, or vice versa. The three time slots 1, 2 and 3 and thesignal sequence SS conform to the American mobile telephone systemstandard TIA/IS-54. In this system, the time slots have a time durationof 6.7 milliseconds, which in the case of the majority of situationsoccurring in practice require the channel estimation circuit 16 to beadaptive, as mentioned above.

As before mentioned in the introduction, problems occur in channelequalization and symbol estimation in digital signal transmissionsystems whose signal bandwidth B exceeds the system symbol rate R=1/TS.This is the case, for instance, in the abovementioned American mobiletelephone system, whose signal bandwidth is B=30 kHz and whose symbolrate R=24.3 kBd. According to the sampling theorem, it is practicallynever sufficient to sample the baseband signal y(T) at the symbol rate Rin systems such as these. However, the symbol rate can be used for thesignal processing in accordance with the viterbi algorithm in thechannel equalizer 17. This is possible when the baseband signal y(T) issampled at a higher frequency than the symbol rate R and this higherfrequency is used in accordance with the invention when prefiltering inthe prefilter circuit 20 and when generating filter coefficients in thechannel estimation circuit 16. The use of the symbol rate in the channelequalizer 17 allows the equalizer to be relatively simple, and theinvention also enables the simplified non-quadratic metric calculationfor the viterbi algorithm to be used.

The receiver shown schematically on the right half of FIG. 1 isillustrated in more detail in FIG. 4. The radio receiver 14 is connectedto the correlating and sampling unit 15, which includes a first sampler21, a second sampler 22, a correlating circuit 23, a synchronizingcircuit 24 and a generator 25 for the synchronizing sequence SY known tothe receiver. The first sampler 21 receives the continuous basebandsignal y(T) from the radio receiver 14 and samples this signal eighttimes for each symbol, i.e. it has a sampling frequency of 8/TS. Thethus sampled signal, referenced y(k/8), is delivered to the correlatingcircuit 23. A first channel estimate F for the observed symbol sequenceSS is generated in this circuit with the aid of the synchronizingsequence SY from the generator 25 and the transmitted, observedsynchronizing sequence. When generating this first channel estimate,there is also established a symbol sampling time point TO in thesynchronizing circuit 24. This symbol sampling time point controls thesecond sampler 22, by which, in accordance with this example, two of theoriginal eight sampling time points for each symbol are chosen with atime spacing of TS/2. This results in the observed signal y(k/2), whichthe sampling unit delivers to the prefilter circuit 20 and the channelestimation circuit 16. Down-sampling in the unit 22 is effected so as tosimplify signal processing in the aforesaid channel estimation circuit.The original eight samplings are used to establish the symbol samplingtime TO, which is the starting point at which symbols are counted in theaforesaid symbol counter. The symbol sampling time TO and the firstchannel estimate HF are delivered to the channel estimation circuit 16.

A brief description of how the channel estimate F is generated in thecorrelating and sampling circuit 15 will now be given. An impulseresponse which includes the channel 13 is generated with the aid of thesignal y(k/8) and the synchronizing sequence. The impulse responseextends over a time interval which includes several symbol times TS anddiscrete impulse response values are generated at a time spacing ofTS/8. A shorter time interval containing the first channel estimate HFis selected within the aforesaid time interval. In the case of theillustrated embodiment, this choice is made so that the first channelestimate HF obtains maximum energy. Furthermore, the first channelestimate HF is given solely at points which are spaced apart by the timespacing TS/2. The Swedish Patent Application No. 8903842-6 whichcorresponds to U.S. Pat. No. 5,228,057 that is incorporated here byreference, describes in more detail how the channel estimate of maximumenergy is chosen. However, it lies within the scope of the invention tochoose the channel estimate in other known ways. It should be observedthat the channel estimate, both the first channel estimate HF and thelater adapted channel estimate, includes both the physical radio channel13 and the transmitter filter 11 and the receiver filter, for instance aMF-filter. The prefilter circuit 20 is not included in the channelestimate. It should also be noted that by correlation is actually meantgenerally a least square estimation. In principle, an estimation of thisnature coincides with a correlation when the known synchronizingsequence used has a so-called white noise autocorrelation function. Thisis often the case in mobile telephony systems.

The channel estimation circuit 16 includes an adaptive channelestimation filter 31, a delay circuit 32, a difference former 33, acircuit 34 which performs an adaptation algorithm, a quadrating and meanvalue forming circuit 35, a signal switch 36, a symbol generator 37, afilter generator 38 which includes a prediction circuit 38A, and ametric calculation filter 39. The channel estimation filter 31 receivesthe first channel estimate HF and the symbol sampling time point TO, andalso the symbols S_(D) (k) estimated in the channel equalizer 17. Theestimated signal values y(k/2) delivered to the difference former 33 areformed with the aid hereof. The difference former 33 also receives theobserved signal y(k/2) which has been delayed in the circuit 32, anddelivers an error signal e(k/2)=y(k/2)-y(k/2). The error signal isdelivered to circuit 34, which, through its adaptation algorithm,controls the adaptive filter 31. It also delivers successively adaptedvalues H(k/2) of the channel estimate to the filter generator 38, viathe prediction circuit 38A. The filter generator 38 also receivesweighting factors α(k/2)=α_(k-1/2), α_(k) that have been generated inthe circuit 35 with the aid of the error signal e (k/2), as explained inmore detail below. There is generated in the filter generator aprefilter function G(k/2) having filter coefficients which are deliveredto the prefilter circuit 20 and also to the metric calculation filter39. There is generated in this latter filter a filter function W(k)having coefficients for the simplified metric calculation according tothe viterbi algorithm, and the coefficients are delivered to the channelequalizer 17. The channel equalizer 17 receives from the symbolgenerator 37 hypothetical symbols S(k), which assume the four symbolvalues S₀, S₁, S₂ and S₃ given in FIG. 3. The signal switch 36 iscontrolled from the synchronizing circuit 24 and shifts with a spacingof one-half symbol time, TS/2, in alternating an estimated symbol S_(D)(k) and a fictive symbol Ω which has a zero-value. This zero-value shallnot be confused with the binary value 00 of the complexvalue symbolS.sub. O in FIG. 3. The fictive zero-value symbol Ω is in the origin ofthe complex number plane I-Q as shown in FIG. 3. The generating of thefictive symbol Ω has been shown schematically in the Figure byconnecting one terminal 36A of the signal switch 36 to earth potential.The reason why zero-values are shifted in will be explained in moredetail below with reference to FIG. 5.

The prefilter circuit 20 includes a prefilter 26 which receives theprefilter function G(k/2) from the filter generator 38. The observedsignal y(k/2) passes the prefilter and is down-sampled thereafter tosymbol rate in a third sampler 27, which is controlled from thesynchronizing circuit 24. Down-sampling is effected at the symbolsampling time TO, so as to obtain the prefiltered, observed signal z(k),which thus occurs with one value for each symbol time TS.

FIG. 5 illustrates the channel estimation filter 31, the delay circuit32, the difference former 33 and the circuit 34 with the adaptationalgorithm. The filter 31 includes delay circuits 41, coefficientcircuits 42, summators 43 and a switch 44. The delay circuits 41 areconnected sequentially in series and delay the incoming signalsuccessively by one-half symbol time TS/2. The successively delayedsignals are multiplied in the coefficient circuits 42 by coefficients H₀(k), H₁ (k), H₂ (k) and H₃ (k) which are the values of the channelestimate H(k/2) at four time points spaced by one-half symbol time TS/2.The output signals from the coefficient circuits 42 are added in theadders 43 to obtain the estimated signal values y(k/2). The errorsignals e(k/2) are formed in the difference former 33 and delivered tothe adaptation algorithm in the circuit 34. This algorithm is chosen independence on the disturbances that the radio channel 13 is assumed tohave and in the case of the illustrating embodiment is a so-calledLMS-algorithm (Least Mean Square). The output signal from the circuit 34adjusts the coefficients in the coefficient circuits 42, so as tominimize the effect of the error signals e(k/2) in accordance with theLMS-algorithm. The coefficient circuits obtain their starting valuesthrough the first channel estimate HF from the correlating andsynchronizing circuit 15. These starting values are connected with theaid of the switch 44, which is controlled from the synchronizing circuit24. The estimated signal values y(k/2) are generated with the aid of theestimated symbols S_(D) (k), which are delayed through the viterbialgorithm by a number q symbol times TS. The observed signal valuesy(k/2) are therefore delayed the number q symbol times in the delaycircuit 32. By inserting the zero-value fictive symbols Ω between theestimated symbols S_(D) (k), the coefficient circuits 42 obtain azero-value as input signal with each alternate updating. The circuitstherefore need to be updated only once with each symbol time TS, whichsimplifies updating. This will be more apparent from the followingdescription of the channel estimation method.

The estimated signal y(k/2) has two separate values for each symbol,firstly y(k) at the symbol sampling time point TO and secondly y(k-1/2)one-half symbol time TS/2 earlier. These values are generated inaccordance with the following:

    y(k-1/2)=H.sub.o (k) S.sub.D (k)+H.sub.2 (k) S.sub.D       (k-1)

    y(k)=H.sub.1 (k) S.sub.D (k)+H.sub.3 (k) S.sub.D (k-1)     (1)

In FIG. 5, the symbol values of the symbol sequence S_(D) (k),Ω at timeposition k-1/2 one-half symbol time TS/2 prior to the symbol samplingtime TO are marked at the inputs of the coefficient circuits 42.One-half symbol time later, at symbol sampling time TO, the symbolvalues are shifted TS/2 to the right in the Figure. The error signalse(k/2) during a symbol time have two different values during the symboltime TS:

    e(k-1/2)=Y(k-1/2)-y(k-1/2)

    e(k)=y(k)-y(k)                                             (2)

where y (k) and y (k-1/2) are the two observed signal values during asymbol time of the observed signal y(k/2). The channel estimate isupdated in the case of the illustrated embodiment through theLMS-algorithm according to the relationships: ##EQU1## In the above listof relationships, μ is a parameter, the step length, in the adaptationalgorithm. It will be seen from the relationship (3) that the values ofthe coefficient circuits 42 need only be calculated once with eachsymbol time, as a result of inserting the zero-value fictive symbols Ω.It will also be seen from the relationship (1), generation of theestimated signals y(k/2) is also simplified by the insertion of thezero-value fictive symbol Ω. Each of the relationships (1) has only twoterms instead of the four terms that would be required if values otherthan zero-values were inserted between the estimated symbols S_(D) (k)and S_(D) (k-1). The RLS-algorithm (Recursive Least Square) or theKalman-algorithm are examples of alternatives to the LMS-algorithm.

An example of the possible configuration of the channel estimate isshown in FIG. 6, which is a diagram in which the coordinate axes denotewith the time T and the channel estimate amplitude H. A curve A shows acontinuous impulse response for the channel 13 and the discrete valuesH₀ (k), H₁ (k), H₂ (k) and H₃ (k) for the channel estimate are shown atthe selected time points at the time spacing TO. The Figure shows thesymbol sampling time TO, and the symbol counter k indicates that thediscrete channel estimate values relate to the time index k.

The prefilter function G(k/2) can be said generally to be a function ofthe channel estimate (H(k/2) and the weighting factors α(k/2). Theoptimal setting of the prefilter is a filter matched to the radiochannel, provided that the number of coefficients in the channelestimate and the prefilter is sufficiently large. As before mentioned,the use of such long channel estimates is encumbered with drawbacks. Oneof the more significant advantages that can be achieved with the presentinvention is that the channel estimate and the prefilter can be madevery short. This is made possible by the insertion of the weightingfactors α(k/2). According to the illustrated embodiment, the prefilterfunction G(k/2) has four coefficients, which are generated in the filtergenerator 38 according to the relationships: ##EQU2## The coefficientvalues in the channel estimate H(k/2) are complex conjugated asindicated by the symbol *, and arranged in reverse time order, as shownby the order of the indexes 0, 1, 2, 3, and are multiplied by the weightfactors α(k/2). It is assumed in the relationships (4) that theprediction circuit 38A is disconnected, so that the channel estimateH(k/2) is used directly when generating the prefilter function G(k/2).

The filter generator 38 delivers the coefficients in the relationship(4) to the prefilter 26, as shown in FIG. 7. The prefilter includesdelay circuits 261, coefficient circuits 262 and summators 263. Thedelay circuits are connected sequentially in series and delay theincoming, observed signal y(k/2) successively by one-half symbol timeTS/2. The delayed signals are multiplied in the coefficient circuits 262by the coefficient values in accordance with the relationship (4) andare added in the adders 263. The resultant summation signal is sampledonce for each symbol time TS, so as to obtain the prefiltered signalz(k).

According to one alternative, the prediction circuit 38A is used so thatthe predicted values of the channel estimate are utilized whengenerating the prefilter function G(k/2). This is significant wheneffecting the metric calculation in the channel equalizer 17, where thechannel estimate coefficients vary with time. As will be apparent fromthe relationships 4, the coefficients in the channel estimate H(k/2) areused in the reverse order when generating the prefilter function G(k/2).Thus, in the case of the illustrated embodiment, a signal value of thesignal y(k/2) at time point k will be multiplied by a coefficient valuein the first coefficient of the prefilter that applies for the timepoint k-1. This can result in some impairment of the symbol estimationin the case of highly pronounced time dispersion of the channel 13. Thisis counteracted by predicting with prediction times of the prefiltercoefficients of mutually different durations, the longest predictiontime for the first coefficient of the prefilter and successivelydecreasing prediction times for the following coefficients. Predictionin the circuit 38A results in predicted coefficient values in theprefilter function G(k/2) which are instantaneous with the signal valuesy(k/2). Reference is made to a dissertation submitted to the Kungl.Tekniska Hogskolan (The Royal Institute of Technology) in Stockholm byErik Dahlman: "A Study of Adaptive Detectors for Fast-VaryingMobile-Radio Channels", Oct. 19, 1992, Report No. TRITA-TTT-9102,Section 3, "Improved Channel Estimates Using Prediction", for a morecomprehensive description of prediction methods.

The coefficients of the filter function W(k) are used for performing themetric calculation in the channel equalizer 17, these coefficients beinggenerated in the metric calculation filter 39. The filter function isgenerated as a convolution of the channel estimate H(k/2) with theprefilter function G(k/2), H(k/2) x G(k/2), where the symbol x signifiesthe convoluting operation. The result of this convolution is sampled atsymbol rate with a starting point from the symbol sampling time point,as indicated by the reference TO in FIG. 4. This sampling is effected ina known manner, with the aid of a sampling unit not shown.

The channel equalizer 17 works in accordance with a so-called symbolsampled viterbi algorithm, since it channel-equalizes the signal z(k)which is sampled at symbol rate. Reference is made to the aforesaidreference "The Viterbi Algorithm" by G. Forney for a more detaileddescription of the viterbi algorithm. The algorithm has, in a knownmanner, a number of states N=M^(L), where M signifies the number ofvalues that a symbol may have and L is the memory length of the filterfunction W(k) in the number of symbol times TS. In the illustratedembodiment, M=4 according to FIG. 3 and, after convolution, the memorylength of the filter function W(k) has been chosen as L=1, so that theequalizer 17 will have N=4 number of states. It can be said generallythat the viterbi algorithm compares sequences of the observed,prefiltered signals z(k) with hypothetical sequences that are generatedwith the aid of the hypothetical symbols S(k) and with the aid of thecoefficients of the filter function W(k). The hypothetical symbols aregiven by the set:

    S(k)=(S.sub.0 (k), S.sub.1 (k), S.sub.2 (k), S.sub.3 (k) ) (5)

The comparison between the two sequences results in deviation values,called metric values, which are calculated stepwise, by adding the deltametric values. As before mentioned, the generation of delta metricvalues in accordance with the present invention is a simplifiednon-quadratic process and is carried out in the manner described in theaforesaid reference "Adaptive Maximum-Likelihood Receiver forCarrier-Modulated Data-Transmission Systems" by G. Ungerboeck. The deltametric values are calculated for different transitions between thestates of the viterbi algorithm. The largest metric value is chosen foreach step in the generation of the metric value and correspondingtransitions are noted. Generation of the metric values is interruptedafter a predetermined number of calculation steps and a symbol ischosen, such as the estimated symbol S_(D) (k) on the basis of themetric value obtained. It will also be noted that in the simplifiedviterbi algorithm, the largest metric value is chosen for each state asdistinct from what is the case with typically used variants of theviterbi algorithm. The simplification resulting from the use ofnon-quadratic metric calculations affords important advantages inpractice in symbol estimation, because quadration or squaring of thevalues obtained is avoided.

As before mentioned, the estimated symbols S_(D) (k) are delayed by anumber q symbol times in relation to the received signal y(k/2). Thisresults in a delay in adaptation of the channel estimation filter 31. Inorder to reduce this harmful delay, there is used in accordance with oneinventive alternative preliminary estimated symbols S_(p) (k) from thechannel equalizer 17 in the adaptation process. The preliminary symbolsS_(p) (k) are decided after a fewer number of steps than the finalsymbols S_(D) (k) in the equalizer 17 and are produced with a smallerdelay than the aforesaid q symbol times, as indicated in FIG. 4.

According to one advantageous embodiment of the invention, the aforesaidweight factors α(k/2)=(α_(k-1/2), α_(k)) are generated with the aid ofthe error signals e(k) and e(_(k-1/2)). This generation of the weightfactors is based on the observation that the statistic expected valuesof the respective squares of the error signals represent a total noisedisturbance level, residual intersymbol interference and co-channelinterference. The greater the expected value, the less thecorrespondence between the estimated signal values y(k-1/2) and y(k) andtheir respective observed signals y(k-1/2) and y(k). It follows fromthis that also the corresponding delta metric values will correspondless accurately to the error caused by noise on the channel. These deltametric values shall then be weighted down in the viterbi algorithm,since the values are uncertain. It may be that one expected value, oractually the statistic variance, is ten times greater than the other,particularly in the case of short channel estimates. The correspondingerror signal will then probably contain a large amount of unknownintersymbol interference rather than channel noise and is thereforeuncertain. The powers of the two error signals e(k) and e(k-1/2) willdiffer greatly from one another, as before mentioned, particularly whenthe channel estimate H(k/2) has few coefficients. A good symbolestimation can be made by utilizing the weight factors α(k/2), even whenthe channel estimate has few coefficients. This is one of thefundamental advantages afforded by the present invention. It will beunderstood, however, that embodiments which include relatively longchannel estimates also lie within the scope of the invention. In thiscase, symbol estimation requires more calculation, but generation of theweighting factors is simplified and these factors can be adjusted toα(k/2)=1, or they could at least be made the same for all taps. In thecase of these latter embodiments of the invention, it remains togenerate the prefilter with the aid of the channel estimate, so as toavoid the problems of stability and quickness in symbol estimation,mentioned in the introduction.

The statistical expected values are estimated by squaring the values ofthe error signals and forming mean values. The expected values and theweight factors are generated in the circuit 35, which is shown in moredetail in FIG. 8. The circuit includes two quadrators 51 and 52, twolowpass filters 53 and 54, two inverters 55 and 56 and two signalswitches 57 and 58. The signal switch 57 receives the error signalse(k/2) and delivers these signals alternately to the quadrators 51 and52 at one-half symbol time TS/2 intervals. The signal switch 57 iscontrolled in a manner not more closely shown by signals from thesynchronizing circuit 24 in FIG. 4. The two error signals e(k-1/2) ande(k) are squared in their respective quadrators 51 and 52 and meanvalues are formed from the squared values by filtering said values intheir respective lowpass filter 53 and 54. These filters deliver signalsσ² (k-1/2) and σ² (k) respectively which correspond to the aforesaidstatistical expected values of the error signals. The signals σ² (k-1/2)and σ² (k) are inverted in respective inverters 55 and 56 to provide theaforesaid weight factors α_(k) and α_(k-1/2) and are delivered to thesignal switch 58. The signal switch is controlled from the synchronizingcircuit 24, in a manner not shown in detail, and applies the weightfactors to the filter generator 38 at intervals of one-half symbol timeTS/2. The circuit 35 shown in FIG. 4 thus generates weight factors inaccordance with the following relationship:

    α.sub.k-1/2=1/| e(k-1/2)|.sup.2

    αk=1/|e(k) |.sup.2                 (6)

where the horizontal lines above |e(k-1/2)| and |e(k)| the formation ofmean values.

According to one alternative, attention is also paid to the values ofthe filter coefficients in the channel estimate when generating theweight factors according to the following relationship:

    α.sub.k-1/2 =(H.sub.0.sup.2)+H.sub.2.sup.2)/σ.sup.2 (k-1/2)

    α.sub.k =(H.sub.1.sup.2 +H.sub.3.sup.2)/σ.sup.2 (k)(7)

In order to generate these alternative weight factors, the circuit 35receives the channel estimate H(k/2) from the channel estimation circuit31, via a connection 35A shown in broken lines in FIG. 4. The filtercoefficients H₀ and H₂ and H₁ and H₃ are squared and added in pairs inthe circuits 59 and 60, which are included in the quadrating and meanvalue-forming circuit 35, and are multiplied by the inverted values ofσ² (k-1/2) and σ² (k) respectively. The thus generated weight factorsα(k/2) are delivered to the filter generator 38 and used in generatingthe prefilter function G(k/2), as described above with reference to therelationships (4). According to another alternative method of generatingthe weight factors, there is used a more direct measurement of thereceived signal strength in the numerator of the relationships (7). Thismeasurement is the so-called RSSI-value (Received Signal StrengthIndicator) which is mainly a squared and lowpass-filtered value of theabsolute value of the received signal y(k/2). This RSSI-value isgenerated in a circuit 35B, shown in broken lines in FIG. 4.

The embodiment above comprises adapting the channel estimate H(k/2) withthe aid of the decided or preliminarily decided symbols. According to asimplified alternative, the values of the filter coefficients areadjusted in the channel estimation circuit 31 only once with each symbolsequence SS, with the aid of the first channel estimate HF. This meansthat the circuit 34 with the adaptation algorithm is excluded. In thisrespect, the symbol sequence containing alternately estimated symbolsS_(D) (k) and fictive zero-value symbols Ω delivered to the channelestimation circuit 31 are used solely to generate the estimated signaly(k/2). However, according to the simplified alternative, the insertionof the zero-value fictive symbols Ω are significant to the generation oferror signals e(k/2), which according to the relationships (1) and (2)are generated with the aid of the estimated signals S_(D) (k).

The inventive method is outlined in the flowsheet presented in FIG. 9.The radio signal R(T) is received in a block 70, mixed down and filteredto a baseband signal y(T). This signal is sampled eight times per symboltime TS according to block 71, and the sampled signal y(k/8) is used forcorrelation, block 72. This correlation gives the sampled impulseresponse of the channel, including the radio channel 13, which is usedto determine the channel estimate HF and to determine the symbolsampling time TO. The once sampled signal y(k/8) is down-sampled on thebasis of this time point TO according to block 73, to obtain theobserved signal y(k/2), which has two signal values per symbol time TS.The signal y(k/2) is prefiltered in block 74 and down-sampled to symbolrate, so as to obtain the prefiltered signal z(k). According to block75, the estimated symbols S_(D) (k) are decided in the channelequalizer, and according to block 76, the symbol sequence of theseestimated symbols and the fictive zero-value symbols Ω is generated. Theestimated signal values y(k/2) are generated in block 77 with the aid ofthe channel estimate HF and the aforesaid signal sequence. The errorsignals e(k/2) are generated with the aid of these estimated signalvalues and the observed signal values y(k/2) according to block 78. Theweight factors α(k/2) are generated in block 79 by squaring, lowpassfiltering and inverting the error signals. According to block 80, theweight factors α(k/2) and the channel estimate HF are used to generatethe prefilter coefficients. These coefficients are used whenprefiltering according to block 74 and also to generate the metricfilter according to block 81. The coefficients in the metric filter W(k)are used to decide the symbols S_(D) (k).

The flow schematic in FIG. 9 illustrates a simple embodiment of theinvention, for the sake of clarity in full lines. Also shown in the flowschematic, in broken lines, is the block 84 according to which thechannel estimate (H(k/2) is generated and shows how this channelestimate according to block 82 is adapted with the aid of the errorsignal e(k/2). Prediction of the channel estimate coefficients prior togenerating the prefilter is shown with a block 83.

In the above embodiment, the observed sampled signal y(k/2) has twosignal values per symbol time TS. It lies within the scope of theinvention to choose, for instance, four or still more signal values persymbol time. This requires, however, that the channel estimation filter31 has correspondingly more coefficient circuits 42. According to thisexample, the channel estimate H(k/2) extends over one symbol time, butmay be broader. This also requires the channel estimation filter 31 tohave more coefficient circuits 42, which results in a larger adaptationdelay. The preliminarily decided symbols S_(P) (k) can be used with theintention of reducing the delay when adapting the channel estimateH(k/2) and generating the weight factors α(k/2). It is also fullypossible to have a relatively long channel estimation filter andtherewith a long prefilter without extending the metric calculationfilter, as this filter can be truncated. This enables the number ofstates in the viterbi analyzer 17 that are dependent on the length ofthe metric calculation filter to be limited, despite the channelestimate being long.

An alternative embodiment of the invention will now be described withreference to FIG. 10. The embodiment comprises essentially a receiverhaving antenna diversity, in which the received signal R(T) is processedas described above in two separate receiver circuits 101 and 102, whichform two separate diversity branches. Output signals from these circuitsare combined and processed in the channel equalizer 17. Each of the twocircuits 101 and 102 includes a respective radio receiver 141 and 142which receives the signal R(T) on its respective antenna. Each of theradio receivers delivers a respective baseband signal y1(T) and y2(T) toits respective correlating and sampling circuit 151 and 152. In additionto generating the symbol sampling time point TO, each of these circuitsalso generates a respective first channel estimate HF and H2F, and arespective observed sampled signal y1(k/2) and y2(k/2). These signalsare prefiltered in prefilter circuits 201 and 202 to obtain prefiltered,observed signals z1(k) and z2 (k) respectively. Each of the receivercircuits 101 and 102 has a respective channel estimation circuit 161 and162 which each generate a respective prefilter function G1(k/2) and G2(k/2) and a respective filter function W1(k) and W2 (k) for metriccalculation. Each of the channel estimation circuits 161 and 162receives its respective channel estimate values and sampling timepoints, delivers the prefilter functions G1(k/2) and G2(k/2) to arespective prefilter 201 and 202, and produces the filter functionsW1(k) and W2(k). These are combined in a circuit 104 to form the filterfunction W(k) and the prefilter, observed signals z1(k) and z2(k) arecombined in a circuit 103 to form the common, prefiltered signal z(k).Combination of both the filter functions W1(k) and W2(k) and theprefiltered, observed signals z1(k) and z2(k) can be effected byaddition. The channel equalizer 17 generates the estimated symbols S_(D)(k) with the aid of the prefiltered signal z(k) and the filter functionW(k) through its viterbi algorithm with non-quadratic metriccalculation. The channel estimation circuits 161 and 162 are adaptedwith the aid of the estimated symbols S_(D) (k).

It will be observed that in the case of the embodiment having antennadiversity, the different diversity branches 101 and 102 have, in themajority of cases, different symbol sampling time points, even thoughthese time points have been referenced TO in common. Weight factors canbe calculated in the same manner as that in the embodiment describedwith reference to FIGS. 4-9, these weight factors being generallydifferent in the different diversity branches. When adapting the channelestimate, there can be used a zero sequence of the zero-value symbols Ωand this symbol sequence will be the same for both diversity branches,since only one array of the estimated symbols S_(D) (k) is generated.

Still another alternative embodiment of the invention will be describedwith reference to FIG. 11. This embodiment can be used when the channelvaries rapidly but the signal bandwidth B of the system is beneath thesymbol rate R, so that the problem of "excess bandwidth" does not exist.The radio receiver 14 is connected to a correlating and sampling unit315, which includes a sampling unit 321, correlating circuit 23, thesynchronizing circuit 24 and the generator 25 for the synchronizingsequence SY. The sampling unit 321 receives the baseband signal y(T) andsamples this signal at symbol rate TS/1 and delivers the observedsampled signal y(k) to the correlating circuit 23. This circuitgenerates the first channel estimate HF, which is delivered to thesynchronizing circuit 24, which in turn delivers the channel estimateand the synchronizing time point TO to a channel estimation circuit 316.

The channel estimation circuit 316 includes an adaptive channelestimation filter 331, the delay circuit 32, the difference former 33,the adaptation circuit 34, the symbol generator 37, a filter generator338 with a prediction circuit 338A and a metric calculation filter 339.

The correlating and sampling unit 315 delivers the observed signal y(k)to a prefilter 326, which delivers the observed, prefiltered signal z(k)to the channel equalizer 27. The channel equalizer produces the desiredestimated symbols S_(D) (k).

The channel estimation filter 331 receives the first channel estimateHF, the symbol sampling time point TO and the estimated symbols S_(D)(k). The estimated signal values y(k) are formed with the aid hereof andare delivered to the difference former 33. The difference former alsoreceives the observed signal y(k) which has been delayed in the circuit32, and produces an error signal e(k). The error signal is received bythe adaptation circuit 34 which controls the channel estimation filter331 through its algorithm, said filter delivering successively adaptedvalues of the channel estimate H(k) to the filter generator 338, via theprediction circuit 338A. There is generated in the filter generator aprefilter function G(k), whose coefficients are delivered to theprefilter 326 and to the metric calculation filter 339. This filter alsoreceives the channel estimate H(k) and generates through a convolutingoperation the filter function W(k) whose coefficients are delivered tothe channel equalizer 17.

Similar to preceding embodiments, the prediction circuit 338A can beexcluded so as to simplify signal processing. The preliminarilyestimated symbols S_(P) (k) can be used when generating the estimatedsignal values y(k).

When seen against the background of the description of the embodimentsillustrated in FIGS. 1 to 7, the person skilled in this art will readilyunderstand the signal processing that takes place in the channelestimation circuit 316. This signal processing will not therefore bedescribed in detail, and only those essential differences in relation tothe preceding embodiment will be mentioned. In the FIG. 11 embodiment,signal processing takes place at symbol rate and the prefiltered signalis therefore not sampled down. The coefficients of the channelestimation filter 331 are mutually spaced by one symbol time and nocorrespondence to the fictive symbols Ω is generated. Only one errorsignal e(k) per symbol is generated and consequently no error signalweighting factors are generated. The coefficients of the filtergenerator 338 and the metric calculation filter 339 are mutually spacedat a distance of one symbol time. Similar to preceding embodiments, acharacteristic feature is that the system impulse response for thechannel, including transmitter filter and receiver filter, is estimatedexplicitly and that the prefilter and the metric calculation filter arethereafter generated on the basis of the channel estimate.

The above exemplifying embodiments of the invention have been describedwith reference to a time-shared radio communications system according toFIG. 2. However, it lies within the scope of the present invention forthe signals to be transmitted in other formats than the described signalsequences SS. Thus, the invention can be applied to e.g. a solelyfrequency-divided system, in which one or more synchronizing sequencesare transmitted together with the information to be conveyed.

In addition to radio communication, the invention can also be appliedwith digital communication within a network having fixed lines. Twosubscribers can be connected together on different occasions throughchannels which include separate lines having different impulseresponses. Similar to the aforedescribed radio communication embodiment,the receivers of respective subscribers are provided with a correlatingcircuit, prefilter circuit, channel equalizer, and channel estimationcircuit of the same kind as that shown in FIG. 4. Correlation iseffected with the aid of a transmitted synchronizing sequence and thereis formed a channel estimate which is utilized to generate a prefilterand a metric calculation filter. The receivers are automaticallyadjusted to each line, by using the channel equalizer. The invention canthus be applied in every transmission system that uses an equalizer inits receiver.

As before mentioned, adaptation of the channel estimation filter 31 issimplified by inserting the fictive zero-value symbols Ω, and theadaptation delay is relatively small. In known techniques, for instancethe technique according to the aforesaid article in IEEE by YongbingWan, et al, there are used interpolated symbol values between theestimated symbol values, such as fictive symbols. This results in adelay when adapting the channel estimate, which always impairs the finalsymbol estimation. One drawback with the technique disclosed in thearticle is that the filters in the transmission chain, transmitter andreceiver filters, must be known to a high degree of accuracy, which isnot the case when the zero-value symbols (Ω) are used. Another drawbackis that the complexity of the receiver increases. The inventivegeneration of the prefilter function G(k/2) with the aid of the channelestimate H(k/2) and the weight factors α(k/2) provides importantadvantages. The simplified, non-quadratic metric calculation can be usedadvantageously in the viterbi analysis in the channel equalizer 17,which is highly significant to effecting symbol estimation in practice.The channel estimate can be allowed to have relatively few coefficientsand the viterbi algorithm in the equalizer 17 has a correspondinglysmall number of states. In the case of fast varying channels, forinstance with fading, the channel estimate H(k/2), the prefilter G(k/2)and the metric calculation filter W(k/2) remain stable without needingto introduce harmful restrictions on the adaptation of the channelestimate.

What is claimed is:
 1. In a method of receiving successive digitalsymbols transmitted over a communication channel that uses a viterbialgorithm, a method of estimating transmitted symbols comprising thesteps of:(a) sampling a received signal at a sampling time point duringeach received symbol; (b) correlating the samples generated by step (a)with a predetermined sequence to produce at least an initial estimatedimpulse response of the communication channel; (c) generatingcoefficients of a prefilter based on an adapted estimated impulseresponse and prefiltering the samples; (d) generating coefficients of ametric calculation filter based on the coefficients of the prefilter andthe adapted estimated impulse response; (e) generating at leastpreliminary estimated symbols using the viterbi algorithm and theprefiltered samples and the coefficients of the metric calculationfilter; (f) generating estimated signal values based on the estimateimpulse response and the at least preliminary estimated symbols; (g)generating error signal according to differences between the samplesgenerated by step (a) and the estimate signal values; and (h) generatingthe adapted estimated impulse response based on the at least preliminaryestimated symbols and the error signals.
 2. The method of claim 1,wherein step (d) includes the step of convolving the estimated impulseresponse with the coefficients of the prefilter.
 3. The method of claim1, wherein step (e) includes the following steps:(a) sampling theprefiltered samples at symbol rate; (b) sampling the coefficients of themetric calculation filter at symbol rate; and (c) performing thepreliminary symbol estimation at symbol rate.
 4. In a method ofreceiving successive digital symbols transmitted over a communicationchannel that uses a viterbi algorithm, a method of estimatingtransmitted symbols comprising the steps of:(a) sampling a receivedsignal at a recurrent plurality of time points during each receivedsymbol; (b) designating one of the plurality of time points as a symbolsampling time point; (c) selecting at least two samples for eachreceived symbol, the selected samples being generated at the symbolsampling point and at least one other time point; (d) correlating thesamples generated by step (a) with a predetermined sequence to produceat least an initial estimated impulse response of the communicationchannel; (e) generating coefficients of a prefilter based on theestimated impulse response and weight factor corresponding to the symbolsampling time point and the at least one other time point, andprefiltering the selected samples; (f) generating coefficients of ametric calculation filter based on the estimated impulse response andthe coefficients of the prefilter; and (g) generating at leastpreliminary estimated symbols according to the viterbi algorithm usingthe prefiltered samples and the coefficients of the metric calculationfilter.
 5. The method of claim 4, wherein the step of generatingcoefficients of a prefilter based on the estimated impulse response andthe weight factors includes the steps of:(a) generating complexconjugated values of the estimated impulse response; (b) reordering thecomplex conjugated values in a reverse time order; and (c) multiplyingthe reordered complex conjugated values by the weight factors.
 6. Themethod of claim 4, further comprising the steps of:(a) generatingestimated signal values corresponding to the symbol sampling time pointsand the other time points based on the estimated impulse response andthe at least preliminary estimated symbols and intermediate fictivesymbols; and (b) generating error signals according to differencesbetween the selected samples of the received signal and the estimatedsignal values, wherein the weight factors are generated based on theerror signals.
 7. The method of claim 6, wherein the estimated impulseresponse of the channel is adapted at least once using the preliminaryestimated symbols and the error signals, according to a selectedadaptation algorithm.
 8. The method of claim 6, wherein the weightfactors are generated by squaring, lowpass filtering and inverting theerror signals.
 9. The method of claim 8, wherein the weight factors aregenerated by squaring and lowpass filtering the selected samples andmultiplying the squared, lowpass-filtered selected samples by thesquared, lowpass-filtered and inverted error signals.
 10. The method ofclaim 8, wherein the weight factors are generated by squaring and addingcoefficient values of the estimated impulse response and multiplying thesquared, added coefficient values of the estimated impulse response andthe squared, lowpass-filtered and inverted error signals to form theweight factors.
 11. The method of claim 6, wherein the intermediatefictive symbols are zero-value symbols.
 12. The method of claim 4,further comprising a step of adapting the estimated impulse responseby:(a) generating estimated signal values corresponding to the symbolsampling time points and the other time points based on the estimatedimpulse response and the at least preliminary estimated symbols andintermediate fictive symbols; (b) generating error signals according todifference between the selected samples of the received signal and theestimated signal values; and (c) adapting the estimated impulse responseusing the error signals, the preliminary estimated symbols, and aselected adaptation algorithm.
 13. In a receiver of successive digitalsymbols transmitted over a communication channel that uses a viterbialgorithm, an apparatus for estimating transmitted symbolscomprising:means for sampling a received signal at a sampling time pointduring during each received symbol; means for correlating the samplesgenerated by the sampling means with a predetermined sequence to produceat least an initial estimated impulse response of the communicationchannel; means for generating prefilter coefficients based on an adaptedestimated impulse response and prefiltering the samples; a metriccalculation filter having coefficients based on the prefiltercoefficients and the adapted estimated impulse response; and channelequalizer means for generating at least preliminary estimated symbolsusing the viterbi algorithm based on the prefiltered samples and thecoefficients of the metric calculation filter; means for generatingestimated signal values based on the estimated impulse response and thepreliminary estimated symbols; means for generating error signalaccording to differences between the samples of the received signal andthe estimated signal values; and means for generating the adaptedestimated impulse response using the preliminary estimated symbols andthe error signal, according to a chosen adaptation algorithm.
 14. Thearrangement of claim 13, wherein the metric calculation filter includesmeans for convolving the estimated impulse response with the prefiltercoefficients.
 15. In a receiver of successive digital symbolstransmitted over a communication channel that uses a viterbi algorithm,an apparatus for estimating transmitted symbols comprising:means forsampling a received signal at a recurrent plurality of time pointsduring each received symbol; means for designating one of the pluralityof time points as a symbol sampling time point; means, connected to thesampling means, for selecting at least two samples for each receivedsymbol, the selected samples being the samples generated at the symbolsampling time point and at least one other time point; means forcorrelating the samples generated by the sampling means with apredetermined sequence to produce at least an initial estimated impulseresponse of the communication channel; means for generating coefficientsof a prefilter based on the estimated impulse response and weightfactors corresponding to the symbol sampling time point and the at leastone other time point, and prefiltering the samples; a metric calculationfilter having coefficients based on the prefilter coefficients; channelequalizer means for generating at least preliminary estimated symbolsaccording to the viterbi algorithm based on the prefiltered samples andthe coefficients of the metric calculation filter; and means forgenerating the weight factor for the corresponding selected samples. 16.The apparatus of claim 15, wherein the prefiltering means includes meansfor sampling prefiltered samples at symbol rate, and the metriccalculation filter includes means for sampling the coefficients of themetric calculation filter at symbol rate.
 17. The apparatus of claim 15,further comprising:means for generating estimated signal valuescorresponding to the symbol sampling time points and the other timepoints based on the estimated impulse response and the at leastpreliminary estimated symbols and intermediate fictive symbols; andmeans for generating error signals according to difference between theselected samples of the received signal and the estimated signal values,wherein the weight factors are generated based on the error signals. 18.The apparatus of claim 17, further comprising means for successivelyadapting the estimated impulse response using the preliminary estimatedsymbols and the error signals, according to a chosen adaptationalgorithm.
 19. The apparatus of claim 17, wherein the weight factorgenerating means further comprises means for squaring, lowpassfiltering, and inverting the error signals to produce the weightfactors.
 20. The apparatus of claim 17, further comprising means forsquaring and adding coefficient values of the estimated impulse responseand multiplying the squared, added coefficient values by the squared,lowpass-filtered and inverted error signals to produce the weightfactors.
 21. The apparatus of claim 15, further comprising means foradapting the estimated impulse response comprising:means for generatingestimated signal values corresponding to the symbol sampling time pointsand the other time points based on the estimated impulse response andthe at least preliminary estimated symbols and intermediate fictivesymbols; means for generating error signal according to differencebetween the selected samples of the received signal and the estimatedsignal values; and means for adapting the estimated impulse responseusing the error signals, preliminary estimated symbols and a selectedadaptation algorithm.
 22. The apparatus of claim 21, wherein theintermediate fictive symbols are zero-value symbols.